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rev. a information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a ad7878 one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 617/329-4700 world wide web site: http://www.analog.com fax: 617/326-8703 ? analog devices, inc., 1997 lc 2 mos complete 12-bit 100 khz sampling adc with dsp interface functional block diagram features complete adc with dsp interface, comprising: track/hold amplifier with 2 m s acquisition time 7 m s a/d converter 3 v zener reference 8-word fifo and interface logic 72 db snr at 10 khz input frequency interfaces to high speed dsp processors, e.g., adsp-2100, tms32010, tms32020 41 ns max data access time low power, 60 mw typ applications digital signal processing speech recognition and synthesis spectrum analysis high speed modems dsp servo control general description the ad7878 is a fast, complete, 12-bit a/d converter with a versatile dsp interface consisting of an 8-word, first-in, first-out (fifo) memory and associated control logic. the fifo memory allows up to eight samples to be digitized before the microprocessor is required to service the a/d con- verter. the eight words can then be read out of the fifo at maximum microprocessor speed. a fast data access time of 41 ns allows direct interfacing to dsp processors and high speed 16-bit microprocessors. an on-chip status/control register allows the user to program the effective length of the fifo and contains the fifo out of range, fifo empty and fifo word count information. the analog input of the ad7878 has a bipolar range of 3 v. the ad7878 can convert full power signals up to 50 khz and is fully specified for dynamic parameters such as signal-to-noise ratio and harmonic distortion. the ad7878 is fabricated in linear compatible cmos (lc 2 mos), an advanced, mixed technology process that com- bines precision bipolar circuits with low power cmos logic. the part is available in four package styles, 28-pin plastic and hermetic dual-in-line package (dip), leadless ceramic chip carrier (lccc) or plastic leaded chip carrier (plcc). product highlights 1. complete a/d function with dsp interface the ad7878 provides the complete function for digitizing ac signals to 12-bit accuracy. the part features an on-chip track/hold, on-chip reference and 12-bit a/d converter. the additional feature of an 8-word fifo reduces the high soft- ware overheads associated with servicing interrupts in dsp processors. 2. dynamic specifications for dsp users the ad7878 is fully specified and tested for ac parameters, including signal-to-noise ratio, harmonic distortion and intermodulation distortion. key digital timing parameters are also tested and specified over the full operating tempera- ture range. 3. fast microprocessor interface data access time of 41 ns is the fastest ever achieved in a monolithic a/d converter, and makes the ad7878 compat- ible with all modern 16-bit microprocessors and digital signal processors.
ad7878Cspecifications C2C rev. a (v dd = +5 v 6 5%, v cc = +5 v 6 5%, v ss = C5 v 6 5%, agnd = dgnd = 0 v, f clk = 8 mhz. all specifications t min to t max , unless otherwise noted.) j, a k, l, b s parameter versions 1 versions version units test conditions/comments dynamic performance 2 signal-to-noise ratio (snr) 3 @ 25 c 70 72 70 db min v in = 10 khz sine wave, f sample = 100 khz t min to t max 70 71 70 db min typically 71.5 db for 0 < v in < 50 khz total harmonic distortion (thd) C80 C80 C78 db max v in = 10 khz sine wave, f sample = 100 khz typically C86 db for 0 < v in < 50 khz peak harmonic or spurious noise C80 C80 C78 db max v in = 10 khz, f sample = 100 khz typically C86 db for 0 < v in < 50 khz intermodulation distortion (imd) second order terms C80 C80 C78 db max fa = 9 khz, fb = 9.5 khz, f sample = 50 khz third order terms C80 C80 C78 db max fa = 9 khz, fb = 9.5 khz, f sample = 50 khz track/hold acquisition time 2 2 2 m s max see throughput rate section dc accuracy resolution 12 12 12 bits minimum resolution for which no missing codes are guaranteed 12 12 12 bits relative accuracy 1/2 1/4 1/2 lsb typ differential nonlinearity 1/2 1/2 1/2 lsb typ bipolar zero error 6 6 6 lsb max positive full-scale error 4 6 6 6 lsb max negative full-scale error 4 6 6 6 lsb max analog input input voltage range 3 3 3 volts input current 550 550 550 m a max reference output 5 ref out 3 3 3 v nom ref out error @ 25 c 10 10 10 mv max t min to t max 15 15 15 mv max reference load sensitivity ( d ref out/ d i) 1 1 1 mv max reference load current change (0 m aC500 m a). reference load should not be changed during conversion logic inputs input high voltage, v inh +2.4 +2.4 +2.4 v min v cc = +5 v 5% input low voltage, v inl +0.8 +0.8 +0.8 v max v cc = +5 v 5% input current, i in 10 10 10 m a max v in = 0 to v cc input capacitance, c in 6 10 10 10 pf max logic outputs output high voltage, v oh +2.7 +2.7 +2.7 v min i source 40 m a output low voltage, v ol +0.4 +0.4 +0.4 v max i sink = 1.6 ma db11Cdb0 floating state leakage current 10 10 10 10 m a max floating state output capacitance 6 15 15 15 15 pf max conversion time 7/7.125 7/7.125 7/7.125 m s min/ m s max assuming no external read/write operations 7/9.250 7/9.250 7/9.250 m s min/ m s max assuming 17 external read/write operations see internal comparator timing section power requirements v dd +5 +5 +5 v nom 5% for specified performance v cc +5 +5 +5 v nom 5% for specified performance v ss C5 C5 C5 v nom 5% for specified performance i dd 13 13 13 ma max cs = dmwr = dmrd = 5 v i cc 100 100 100 m a max cs = dmwr = dmrd = 5 v i ss 6 6 6 ma max cs = dmwr = dmrd = 5 v power dissipation 95.5 95.5 95.5 mw max typically 60 mw notes 1 temperature range as follows: j, k, l versions: 0 c to +70 c; a, b versions: C25 c to +85 c; s version: C55 c to +125 c. 2 v in = 3 v. see dynamic specifications section. 3 snr calculation includes distortion and noise components. 4 measured with respect to the internal reference. 5 for capacitive loads greater than 50 pf a series resistor is required (see internal reference section). 6 sample tested @ +25 c to ensure compliance. specifications subject to change without notice.
ad7878 C3C rev. a limit at t min , t max limit at t min , t max limit at t min , t max parameter (l grade) (j, k, a, b grades) (s grade) units conditions/comments t l 65 65 75 ns max clk in to busy low propagation delay t 2 65 65 75 ns max clk in to busy high propagation delay t 3 2 clk in cycles 2 clk in cycles 2 clk in cycles min convst pulse width t 4 0 0 0 ns min cs to dmrd /register enable setup time t 5 0 0 0 ns min cs to dmrd / register enable hold time t 6 45 60 60 ns min dmrd pulse width 50 50 50 m s max t 7 16 16 16 ns min add0 to dmrd /register enable setup time t 8 0 0 0 ns min add0 to dmrd /register enable hold time t 9 2 41 57 57 ns min data access time after dmrd t 10 3 5 5 5 ns min bus relinquish time 45 45 45 ns max t 11 42 42 55 ns min register enable pulse width 50 50 50 m s max t 12 20 20 30 ns min data valid to register enable setup time t 13 10 10 10 ns min data hold time after register enable t 14 2 41 57 57 ns min data access time after busy t reset 2 clk in cycles 2 clk in cycles 2 clk in cycles min reset pulse width notes 1 timing specifications in bold print are 100% production tested. all other times are sample tested at +25 c to ensure compliance. all input signals are specified with tr = tf = 5 ns (10% to 90% of 5 v) and timed from a voltage level of 1.6 v. 2 t 9 and t 14 are measured with the load circuits of figure 1 and defined as the time required for an output to cross 0.8 v or 2.4 v. 3 t 10 is defined as the time required for the data lines to change 0.5 v when loaded with the circuits of figure 2. specifications subject to change without notice. (v dd = 5 v 6 5%, v cc = 5 v 6 5%, v ss = C5 v 6 5%) timing characteristics 1 figure 1. load circuits for access time figure 2. load circuits for output float delay absolute maximum ratings* (t a = +25 c unless otherwise stated) v dd to dgnd . . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v v cc to dgnd . . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v v ss to dgnd . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 v to C7 v v dd to v cc . . . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +0.3 v agnd to dgnd . . . . . . . . . . . . . . . . . C0.3 v to v dd +0.3 v v in to agnd . . . . . . . . . . . . . . . . . . . . . . . . . C15 v to +15 v ref out to agnd . . . . . . . . . . . . . . . . . . . . . . . . . 0 to v dd digital inputs to dgnd clk in, dmwr , dmrd , reset , cs , convst , add0 . . . . . . . . . . . . C0.3 v to v dd +0.3 v di gital outputs to dgnd alfl , busy . . . . . . . . . . . . . . . . . . C0.3 v to v dd +0.3 v data pins db11Cdb0 . . . . . . . . . . . . . . . . . . . . C0.3 v to v dd +0.3 v operating temperature range j, k, l versions . . . . . . . . . . . . . . . . . . . . . . . 0 c to +70 c a, b versions . . . . . . . . . . . . . . . . . . . . . . . C25 c to +85 c s version . . . . . . . . . . . . . . . . . . . . . . . . . C55 c to +125 c storage temperature range . . . . . . . . . . . . C65 c to +150 c lead temperature (soldering, 10 sec) . . . . . . . . . . . . +300 c power dissipation (any package) to +75 c . . . . . . 1000 mw derates above +75 c by . . . . . . . . . . . . . . . . . . 10 mw/ c *stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. these are stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability warning! esd sensitive device caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the ad7878 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. a. high-z to v oh b. high-z to v ol a. v oh to high-z b. v ol to high-z
ad7878 C4C rev. a pin function description pin pin number mnemonic function 1 1 add0 address input. this control input determines whether the word placed on the output data bus during a read operation is a d ata word from the fifo ram or the contents of the status/control register. a logic low accesses the data word from location 0 of the fifo while a logic high selects the contents of the register (see status/control register section). 1 2 cs chip select. active low logic input. the device is selected when this input is active. 1 3 dmwr dam memory write. active low logic input. dmwr is used in conjunction with cs low and add0 high to write data to the status/control register. corresponds to dmwr (adsp-2100), r/ w (mc68000, tms32020), we (tms32010). 1 4 dmrd data memory read. active low logic input. dmrd is used in conjunction with cs low to enable the three-state output buffers. corresponds directly to dmrd (adsp-2100), den (tms32010). 1 5 busy active low logic output. this output goes low when the adc receives a convst pulse and remains low until the track/hold has gone into its hold mode. the three-state drivers of the ad7878 can be disabled while the busy signal is low (see extended read/write section). this is achieved by writing a logic 0 to db5 ( diso ) of the status/control register. writing a logic 1 to db5 of the status/control register allows data to be accessed from the ad7878 while busy is low. 1 6 alfl fifo almost full. a logic low indicates that the word count (i.e., number of conversion results) in the fifo memory has reached the programmed word count in the status/control register. alfl is updated at the end of each conversion. the alfl output is reset to a logic high when a word is read from the fifo memory and the word count is less than the preprogrammed word count. it can also be set high by writing a logic 1 to db7 ( enaf ) of the status/control register. 1 7 dgnd digital ground. ground reference for digital circuitry. 1 8v cc digital supply voltage, +5 v 5%. positive supply voltage for digital circuitry. 1 9 db11 data bit 11 (msb). three-state ttl output. coding for the data words in fifo ram is twos complement. 10C15 db10Cdb5 data bit 10 to data bit 5. three-state ttl input/outputs. 16C19 db4Cdb1 data bit 4 to data bit 1. three-state ttl outputs. 20 db0 data bit 0 (lsb). three-state ttl output. 21 v dd analog positive supply voltage, +5 v 5%. 22 agnd analog ground. ground reference for track/hold, reference and dac. 23 ref out voltage reference output. the internal 3 v analog reference is provided at this pin. the external load capability of t he reference is 500 m a. 24 v in analog input. analog input range is 3 v. 25 v ss analog negative supply voltage, C5 v 5%. 26 convst convert start. logic input. a low to high transition on this input puts the track/hold into its hold mode and starts conversion . the convst input is asynchronous to clk in and independent of cs , dmwr and dmrd . 27 reset reset. active low logic input. a logic low sets the words in fifo memory to 1000 0000 0000 and resets the alfl output and status/control register. 28 clk in clock input. ttl-compatible logic input. used as the clock source for the a/d converter. the mark-space ratio of this c lock can vary from 35/65 to 65/35. pin configurations lccc plcc dip
ad7878 C5C rev. a ordering guide signal- data temperature to-noise access package model 1, 2 range ratio time options 3 ad7878jn 0 c to +70 c 70 db 57 ns n-28 ad7878aq C25 c to +85 c 70 db 57 ns q-28 ad7878sq C55 c to +125 c 70 db 57 ns q-28 ad7878kn 0 c to +70 c 72 db 57 ns n-28 ad7878bq C25 c to +85 c 72 db 57 ns q-28 AD7878LN 0 c to +70 c 72 db 41 ns n-28 ad7878se 4 C55 c to +125 c 70 db 57 ns e-28a ad7878jp 0 c to +70 c 70 db 57 ns p-28a ad7878kp 0 c to +70 c 72 db 57 ns p-28a ad7878lp 0 c to +70 c 72 db 41 ns p-28a notes 1 to order mil-std-883, class b processed parts, add /883b to part number. contact our local sales office for military data sheet. 2 analog devices reserves the right to ship either ceramic (d-28) packages or cerdip (q-28) hermetic packages. 3 e = leadless ceramic chip carrier; n = plastic dip; p = plastic leaded chip carrier, q = cerdip. 4 available to /883b processing only. status/control register the status/control register serves the dual function of providing control and monitoring the status of the fifo memory. this register is directly accessible through the data bus (db11Cdb0) with a read or write operation while add0 is high. a write operation to the status/control register provides control for the alfl output, bus interface and fifo counter reset. this is normally done on power-up initialization. the fifo memory address pointer is incremented after each conversion and com- pared with a preprogrammed count in the status/control regis- ter. when this preprogrammed count is reached, the alfl output is asserted if the enaf control bit is set to zero. this alfl can be used to interrupt the microprocessor after any predetermined number of conversions (between 1 and 8). the status of the address pointer along with sample overrange and alfl status can be accessed at any time by reading the status/ control register. note: reading the status/control register does not cause any internal data movement in the fifo memory. status information for a particular word should be read from the status register before the data word is read from the fifo memory. status/control register function description db11 ( alfl ) almost full flag, read only. this is the same as pin 6 ( alfl output) status. a logic low indicates that the word count in the fifo memory has reached the preprogrammed count in bit locations db10Cdb8. alfl is updated at the end of conversion. db10Cdb8 (afc2Cafc0) almost full word count, read/write. the count value deter- mines the number of words in the fifo memory, which will cause alfl to be set. when the fifo word count equals the programmed count in these three bits, both the alfl output and db11 of the status register are set to a logic low. for ex- ample, when a code of 011 is written to these bits, alfl is set when location 0 through location 3 of the fifo memory contains valid data. afc2 is the most significant bit of the word count. the count value can be read back if required. db7 ( enaf ) enable almost full, read/write. writing a 1 to this bit disables the alfl output and status register bit db11. db6 (fovr/reset) fifo overrun/reset, read/write. reading a 1 from this bit indicates that at least one sample has been discarded because the fifo memory is full. when the fifo is full (i.e., contains eight words) any further conversion results will be lost. writing a 1 to this bit causes a system reset as per the reset input (pin 27). db5 (foor/diso) fifo out of range/disable outputs, read/write. reading a 1 from this bit indicates that at least one sample in the fifo memory is out of range. writing a 0 to this bit prevents the data bus from becoming active while busy is low, regardless of the state of cs and dmrd . db4 (femp) fifo empty, read only. reading a 1 indicates there are no samples in the fifo memory. when the fifo is empty the internal ripple-down effects of the fifo are disabled and fur- ther reads will continue to access the last valid data word in location 0. db3 (soor) sample out of range, read only. reading a 1 indicates the next sample to be read is out of range, i.e., the sample in location 0 of the fifo. dbCdb0 (fcn2Cfcn0) fifo word count, read only. the value read from these bits indicates the number of samples in the fifo memory. for example, reading 011 from these bits indicates that location 0 through location 3 contains valid data. note: reading all 0s indicates there is either one word or no word in the fifo memory; in this case the fifo empty determines if there is no word in memory. fcn2 is the most significant bit. table i. status/control bit function description bit location db11 db10 db9 db8 db7 db6 db5 db4 db3 db2 db1 db0 status information (read) alfl afc2 afc1 afc0 enaf fovr foor femp soor fcn2 fcn1 fcn0 control function (write) x afc2 afc1 afc0 enaf reset diso xxxxx reset status 1 0 0 0 0 0 0 1 0 0 0 0 x =dont care
ad7878 C6C rev. a internal fifo memory the internal fifo memory of the ad7878 consists of eight memory locations. each word in memory contains 13 bits of information12 bits of data from the conversion result and one additional bit which contains information as to whether the 12- bit result is out of range or not. a block diagram of the ad7878 fifo architecture is shown in figure 3. figure 3. internal fifo architecture the conversion result is gathered in the successive approxima- tion register (sar) during conversion. at the end of conversion this result is transferred to the fifo memory. the fifo ad- dress pointer always points to the top of memory, which is the uppermost location containing valid data. the pointer is incre- mented after each conversion. a read operation from the fifo memory accesses data from the bottom of the fifo, location 0. on completion of the read operation, each data word moves down one location and the address pointer is decremented by one. therefore, each conversion result from the sar enters at the top of memory, propagates down with successive reads until it reaches location 0 from where it can be accessed by a micro- processor read operation. the transfer of information from the sar to the fifo occurs in synchronization with the ad7878 input clock (clk in). the propagation of data words down the fifo is also synchronous with this clock. as a result, a read operation to obtain data from the fifo must also be synchronous with clk in to avoid read/write conflicts in the fifo (i.e., reading from fifo loca- tion 0 while it is being updated). this requires that the micro- processor clock and the ad7878 clk in are derived from the same source. internal comparator timing the adc clock, which is applied to clk in, controls the suc- cessive approximation a/d conversion process. this clock is internally divided by four to yield a bit trial cycle time of 500 ns min (clk in = 8 mhz clock). each bit decision occurs 25 ns after the rising edge of this divided clock. the bit decision is latched by the rising edge of an internal comparator strobe sig- nal. there are 12-bit decisions, as in a normal successive ap- proximation routine, and one extra decision that checks if the input sample is out of range. in a normal successive approxima- tion a/d converter, reading data from the device during conver- sion can upset the conversion in progress. this is due to on-chip transients, generated by charging or discharging the databus, concurrent with a bit decision. the scheme outlined below and shown in figure 4 describes how the ad7878 overcomes this problem. the internal comparator strobe on the ad7878 is gated with both dmrd and dmwr so that if a read or write operation occurs when a bit decision is about to be made, the bit decision point is deferred by one clk in cycle. in other words, if dmrd or dmwr goes low (with cs low) at any time during the clk in low time immediately prior to the comparator strobing edge (t low of figure 4), the bit trial is suspended for a clock cycle. this makes sure that the bit decision is latched at a time when the ad7878 is not attempting to charge or discharge the data bus, thereby ensuring that no spurious transients occur internally near a bit decision point. the decision point slippage mechanism is shown in figure 4 for the msb decision. normally, the msb decision occurs 25 ns after the fourth rising clk in edge after convst goes high. however, in the timing diagram of figure 4, cs and dmrd or dmwr are low in the time period t low prior to the msb deci- sion point on the fourth rising edge. this causes the internal comparator strobe to be slipped to the fifth rising clock edge. the ad7878 will again check during a period t low prior to this fifth rising clock edge; and if the cs and dmrd or dmwr are still low, the bit decision point will be slipped a further clock cycle. the conversion time for the adc normally consists of the 13- bit trials described above and one extra internal clock cycle during which data is written from the sar to the fifo. for an 8 mhz input clock this results in a conversion time of 7 m s. however, the software routine servicing the ad7878 has the potential to read 16 times from the device during conversion8 reads from the fifo and 8 reads from the status/control register. it also has the potential to write once to the status/control register. if these figure 4. operational timing diagram
ad7878 C7C rev. a 17 (16 read plus 1 write) operations all occur during t low time periods, the conversion time will slip by 17 clk in cycles. therefore, if read or write operations can occur during t low periods, it means that the conversion time for the adc can vary from 7 m s to 9.12 m s (assuming 8 mhz clk in). this calcula- tion assumes there is a slippage of one clk in cycle for each read or write operation. initiating a conversion conversion is i nitiated on the ad7878 by ass erting the convst input. this convst input is an asynchronous input indepen- dent of either the adc or dsp clocks. this is essential for applica- tions where precise sampling in time is important. in these applica- tions the signal sampling must occur at exactly equal intervals to minimize errors due to sampling uncertainty or jitter. in these cases the convst input is driven from a tamer or some precise clock source. on receipt of a convst pulse, the ad7878 acknowl- edges by taking the busy output low. this busy output can be used to ensure no bus activity while the track/hold goes from track to hold mode (see extended read/write section). the convst input must stay low for at least two clk in periods. the track/ hold amplifier switches from the track to hold mode on the rising edge of convst and conversion is also initiated at this point. the busy output returns high after the convst input goes high and the adc begins its successive approximation routine. once conversion has been initiated another conversion start should not be attempted until the full conversion cycle has been com pleted. figure 5 shows the taming diagram for the conversion start. in applications where precise sampling is not critical, the convst pulse can be generated from a microprocessor wr or rd line gated with a decoded address (different from the ad7878 cs address). note that the convst pulse width must be a minimum of two ad7878 clk in cycles. figure 5. conversion start timing diagram read/write operations the ad7878 read/write operations consist of reading from the fifo memory and reading and writing from the status/control register. these operations are controlled by the cs , dmrd , dmwr and add0 logic inputs. a description of these operations is given in the following sections. in addition to the basic read/write operations there is an extended read/write operation. this can occur if a read/write operation occurs during a convst pulse. this extended read/write is intended for use with microproces- sors that can be driven into a wait state, and the scheme is recommended for applications where an external timer controls the convst input asynchronously to the microprocessor read/ write operations. basic read operation figure 6 shows the timing diagram for a basic read operation on the ad7878. cs and dmrd going low accesses data from either the status/control register or the fifo memory. a read operation with add0 low accesses data from the fifo while a read with add0 high accesses data from the status/ control register. figure 6. basic read operation basic write operation a basic write operation to the ad7878 status/control register consists of bringing cs and dmwr low with add0 high. in- ternally these signals are gated with clk in to provide an internal register enable signal (see figure 7). the pulse width of this register enable signal is effectively the overlap between the clk in low time and the dmwr pulse. this may result in shorter write pulse widths, data setup times and data hold times than those given by the microprocessor. the timing on the ad7878 timing diagram of figure 8 is there- fore given with respect to the internal register enable signal rather than the dmwr signal. figure 7. dmwr internal logic figure 8. basic write operation
ad7878 C8C rev. a extended read/write operation as described earlier, a read/write operation to the ad7878 can cause spurious on-chip transients. should these transients occur while the track/hold is going from track to hold mode, it may result in an incorrect value of v in being held by the track/hold amplifier. because the convst input has asynchronous capa- bility, a read/write operation could occur while convst is low. the ad7878 allows the read/write operation to occur but has the facility to disable its three-state drivers so that there is no data bus activity and, hence, no transients while the track/ hold goes from track to hold. writing a logic 0 to db5 ( diso ) of the status/control register prevents the output latches from being enabled while the ad7878 busy signal is low. if a microprocessor read/write operation can occur during the busy low time, the busy should be gated with cs of the ad7878 and this gated signal used to stretch the instruction cycle using dmack (adsp- 2100), ready (tms32020) or dtack (68000). when convst goes low, the ad7878 acknowledges it by bringing busy low on the next rising edge of clk in. with a logic 0 in db5, the ad7878 data bus cannot now be enabled. if a read/write operation now occurs, the busy and cs gated signal drives the microprocessor into a wait state, thereby extending the read/write operation. busy goes high on the second rising edge of clk in after convst goes high. the ad7878 data outputs are now enabled and the microprocessor is released from its wait state, allowing it to complete its read/ write operation to the ad7878. the microprocessor cycle time for the read/write operation is extended by the convst pulse width plus two clk in peri- ods worst case. this is the maximum length of time for which busy can be low. assuming a convst pulse width of two clk in periods and an 8 mhz clk in, the instruction cycle is extended by 500 ns maximum. figure 9 shows the timing diagram for an extended read operation. in a similar manner, a write operation will be extended if it occurs during a convst pulse. for processors that cannot be forced into a wait state, writing a logic 1 into db5 of the status/control register allows the out- put latches to be enabled while busy is low. in this case busy still goes low as before, but it would not be used to stretch the read/write cycle and the instruction cycle continues as normal (see figures 6 and 8). figure 9. extended read operation ad7878 dynamic specifications the ad7878 is specified and 100% tested for dynamic perfor- mance specifications rather than for traditional dc speci fications such as integral and differential nonlinearity. these ac specifi- cations provide information on the ad7878s effect on the spec- tral content of the input signal. hence, the parameters for w hich the ad7878 is specified include snr, harmonic distortion, inter- modulation distortion and peak harmonics. these terms are dis- cussed in more detail in the following sections. signal-to-noise ratio (snr) snr is the measured signal-to-noise ratio at the output of the adc. the signal is the rms magnitude of the fundamental. noise is the rms sum of all the nonfundamental signals (excluding dc) up to half the sampling frequency (f s /2). snr is dependent upon the number of quantization levels used in the digitization process; the more levels, the smaller the quantization noise. the theoretical signal-to-noise ratio for a sine wave input is given by snr = (6.02 n + 1.76) db (1) where n is the number of bits. thus for an ideal 12-bit con- verter, snr = 74 db. the output spectrum from the adc is evaluated by applying a sine-wave signal of very low distortion to the v in input, which is sampled at a 100 khz sampling rate. a fast fourier transform (fft) plot is generated from which the snr data can be ob- tained. figure 10 shows a typical 2048 point fft plot of the ad7878kn with an input signal of 25 khz and a sampling frequency of 100 khz. the snr obtained from this graph is 72.6 db. it should be noted that the harmonics are included in the snr calculation. figure 10. ad7878 fft plot effective number of bits the formula given in (1) relates the snr to the number of bits. rewriting the formula, as in (2), it is possible to get a measure of performance expressed in effective number of bits (n). the effective number of bits for a device can be calculated directly from its measured snr. n = snr 1.76 6.02 (2)
ad7878 C9C rev. a figure 11 shows a typical plot of effective number of bits versus frequency for an ad7878kn with a sampling frequency of 100 khz. the effective number of bits typically falls between 11.7 and 11.85 corresponding to snr figures of 72.2 and 73.1 db. figure 11. effective number of bits vs. frequency harmonic distortion harmonic distortion is the ratio of the rms sum of harmonics to the fundamental. for the ad7878, total harmonic distortion (thd) is defined as: thd = 20 log ( v 2 2 + v 3 2 + v 4 2 + v 5 2 + v 6 2 ) v 1 where v 1 is the rms amplitude of the fundamental and v 2 , v 3 , v 4 , v 5 and v 6 are the rms amplitudes of the second to the sixth harmonic. the thd is also derived from the fft plot of the adc output spectrum. intermodulation distortion with inputs consisting of sine waves at two frequencies, fa and fb, any active device with nonlinearities will create distortion products at sum and difference frequencies of mfa + nfb where m, n = 0, 1, 2, 3 . . . . , etc. intermodulation terms are those for which neither m nor n is equal to zero. for example, the second order terms include (fa + fb) and (fa C fb) while the third order terms include (2fa + fb), (2fa C fb), (fa + 2fb) and (fa C 2fb). using the ccif standard, where two input frequencies near the top end of the input bandwidth are used, the second and third order terms are of different significance. the second order terms are usually distanced in frequency from the original sine waves, while the third order terms are usually at a frequency close to the input frequencies. as a result, the second and third order terms are specified separately. the calculation of the intermodulation distortion is as per the thd specification where it is the ratio of the rms sum of the individual distortion products to the rms am- plitude of the fundamental expressed in dbs. intermodulation distortion is calculated using an fft algorithm but, in this case, the input consists of two equal amplitude, low distortion sine waves. figure 12 shows a typical imd plot for the ad7878. peak harmonic or spurious noise peak harmonic or spurious noise is defined as the ratio of the rms value of the next largest component in the adc output spectrum (up to fs/2 and excluding dc) to the rms value of the fundamental. normally, the value of this specification will be determined by the largest harmonic in the spectrum, but for parts where the harmonics are buried in the noise floor the largest peak will be a noise peak. figure 12. ad7878 imd plot histogram plot when a sine wave of a specified frequency is applied to the v in input of the ad7878 and several million samples are taken, it is possible to plot a histogram showing the frequency of occur- rence of each of the 4096 adc codes. if a particular step is wider than the ideal 1 lsb width, then the code associated with that step will accumulate more counts than for the code for an ideal step. likewise, a step narrower than ideal will have fewer counts. missing codes are easily seen in the histogram plot because a missing code means zero counts for a particular code. large spikes in the plot indicate large differential nonlinearity. figure 13 shows a histogram plot for the ad7878kn with a sampling frequency of 100 khz and an input frequency of 25 khz. for a sine-wave input, a perfect adc would produce a cusp probability density function described by the equation: p ( v ) = 1 p ( a 2 v 2 ) where a is the peak amplitude of the sine wave and p ( v ) is the probability of occurrence at a voltage v . the histogram plot of figure 13 corresponds very well with this cusp shape. the ab- sence of large spikes in this plot indicates small dynamic differ- ential nonlinearity (the largest spike in the plot represents less than 1/4 lsb of dnl error). the ad7878 has no missing codes under these conditions since no code records zero counts. figure 13. ad7878 histogram plot
ad7878 C10C rev. a conversion timing the track-and-hold on the ad7878 goes from track-to-hold mode on the rising edge of convst , and the value of v in at this point is the value which will be converted. however, the conversion actually sorts on the next rising edge of clk in after convst goes high. if convst goes high within ap- proximately 30 ns prior to a rising edge of clk in, that clk in edge will not be seen as the first clk in edge of the con- version process, and conversion will not actually start until one clk in cycle later. as a result, the conversion time (from convst to fifo update) will vary by one clock cycle de- pending on the relationship between convst and clk in. a conversion cycle normally consists of 56 clk in cycles (assuming no read/write operations) which corresponds to a 7 as conversion time. if convst goes high within 30 ns prior to a rising edge of clk in, the conversion time will consist of 57 clk in cycles, i.e., 7.125 m s. this effect does not cause track/hold jitter. internal reference the ad7878 has an on-chip temperature compensated buried zener reference (see figure 14) that is factory trimmed to 3 v 1%. internally, it provides both the dac reference and the dc bias required for bipolar operation. the reference output is available (ref out) and is capable of providing up to 500 m a to an external load. figure 14. ad7878 reference circuit the maximum recommended capacitance on ref out for normal operation is 50 pf. if the reference is required for use external to the ad7878, it should be decoupled with a 200 w resistor in series with a parallel combination of a 10 m f tantalum capacitor and a 0.1 m f ceramic capacitor. these decoupling components are required to remove voltage spikes caused by the internal operation of the ad7878. track-and-hold amplifier the track-and-hold amplifier on the analog input of the ad7878 allows the adc to accurately convert an input sine wave of 6 v peak-peak amplitude to 12-bit accuracy. the input bandwidth of the track/hold amplifier is much greater than the nyquist rate of the adc even when operated at its minimum conversion time. the 0.1 db cutoff frequency occurs typically at 500 khz. the track/hold amplifier acquires an input signal to 12-bit accuracy in less than 2 m s. the operation of the track/hold amplifier is transparent to the user. the track/hold amplifier goes from its tracking mode to its hold mode at the start of conversion on the rising edge of convst and returns to track mode at the end of conversion. analog input figure 15 shows the ad7878 analog input. the analog input range is 3 v into an input resistance of typically 15 k w . the designed code transitions occur midway between successive integer lsb values (i.e., 1/2 lsb, 3/2 lsbs, 5/2 lsbs . . . . fsC3/2 lsbs). the output code is 2s complement binary with 1 lsb = fs/4096 = 6 v/4096 = 1.46 mv. the ideal input/ output transfer function is shown in figure 16. figure 15. ad7878 analog input figure 16. input/output transfer function offset and full-scale adjustment in most digital signal processing (dsp) applications offset and full-scale error have little or no effect on system performance. offset error can always be eliminated in the analog domain by ac coupling. full-scale error effect is linear and does not cause problems as long as the input signal is within the full dynamic range of the adc. some applications may require that the input signal span the full analog input dynamic range and, accord- ingly, offset and full-scale error will have to be adjusted to zero. where adjustment is required, offset must be adjusted before full-scale error. this is achieved by trimming the offset of the op amp driving the analog input of the ad7878 while the input voltage is 1/2 lsb below ground. the trim procedure is as follows: apply a voltage of C0.73 mv (C1/2 lsb) at v 1 and adjust the op amp offset voltage until the adc output code flickers between 1111 1111 1111 and 0000 0000 0000. gain error can be adjusted at either the first code transition (adc negative full scale) or the last code transition (adc positive full scale). the trim procedures for both cases are as follows:
ad7878 C11C rev. a positive full-scale adjust apply a voltage of 2.9978 v (fs/2 C 3/2 lsbs) at v 1 . adjust r2 until the adc output code flickers between 0111 1111 1110 and 0111 1111 1111. negative full-scale adjust apply a voltage of C2.9993 v (Cfs/2 + 1/2 lsb) at v l and ad- just r2 until the adc output code flickers between 1000 0000 0000 and 1000 0000 0001. figure 17. ad7878 full-scale adjust circuit microprocessor interfacing the ad7878 high speed bus timing allows direct interfacing to dsp processors. due to the complexity of the ad7878 internal logic, only synchronous interfacing is allowed. this means that the adc clock must be the same as, or a derivative of, the pro- cessor clock. suitable processor interfaces are shown in figures 18 to 21. ad7878Cadsp-2100/tms32010/tms32020 all three interfaces use an external timer for conversion control, allowing the adc to sample the analog input asynchronously to the microprocessor. the ad7878 alfl output interrupts the processor when the fifo preprogrammed word count is reached. the processor then reads the conversion results from the ad7878 internal fifo memory. figure 18. ad7878Cadsp-2700 interface figure 19. ad7878Ctms32020 interface the interfaces to the adsp-2100 and the tms32020 gate the ad7878 cs and the busy to provide a signal which drives the processor into a wait state if a read/write operation to the adc is attempted while the adc track/hold amplifier is going from the track to the hold mode. this avoids digital feedthrough to the analog circuitry. the tms32020 does not have separate rd and wr outputs to drive the ad7878 dmwr and dmrd inputs. these are generated from the processor strb and r/ w outputs with the addition of some logic gates. figure 20. ad7878Ctms32020 interface ad7878Cm cc8000 this interface also uses an external timer for conversion control as described for the previous three interfaces. it is discussed separately because it needs extra logic due to the nature of its interrupts. the mc68000 has eight levels of external interrupt. when interrupting this processor one of these levels (0 to 7) has to be encoded onto the ipl2 C ipl0 inputs. this is achieved with a 74148 encoder in figure 21, (interrupt level 1 is taken for example purposes only). the mc68000 places this inter- rupt level on address bits a3 to a1 at the start of the interrupt service routine. additional logic is used to decode this interrupt level on the address bus and the fc2Cfc0 outputs to generate a vpa signal for the mc68000. this results in an autovectored interrupt, the start address for the service routine must be loaded into the appropriate auto vector location during initial- ization. for further information on the 68000 interrupts con- sult the 68000 users manual .
ad7878 C12C rev. a the mc68000 as and r/ w outputs are used to generate sepa- rate dmwr and dmrd inputs for the ad7878. as with the three interfaces previously described, wait states are inserted if a read/write operation is attempted while the trac k/hold amplifier is going from the track to the hold mode. figure 21. ad7878Cmc68000 interface typical ad7878 microprocessor operating sequence after power-up or reset, the status/control register is initialized by writing to the ad7878. this enables the alfl output if required for a microprocessor interrupt and sets the effective word length of the fifo memory. the processor now executes the main body of the program while waiting for an adc interrupt. this interrupt will occur when the preprogrammed number of samples are collected in the fifo memory. the interrupt ser- vice routine first interrogates db5(foor) of the status/control register to determine if any sample in the fifo memory is out of range. if all data samples are valid, then the program pro- ceeds to read the fifo memory. if, on the other hand, at least one sample is out of range, then an overrange routine is called. there are many actions that can be taken by the out of range routine, the selection of which is application dependent. one option is to ignore all the current samples residing in the fifo memory, reinitialize the status/control register and return to the main body of the program. another option is to check the indi- vidual out of range status of each word in the fifo memory and to discard the invalid ones. the underrange or overrange status of each word can also be determined and the analog input adjusted accordingly before returning to the main program. note: there is no need to check the out-of-range status if the analog input is always assured to be within range. throughput rate the ad7878 has a maximum specified throughput rate (sample rate) of 100 khz. this is a worst-case test condition and specifi- cations apply for reduced sampling rates, provided that nyquist criterion is obeyed. the throughput rate must take into account adc convst pulse width, adc conversion time and the track/hold amplifier acquisition time. the time required for each of these tasks is shown in table ii for a selection of dsp proces- sors. since the adc clock has to be synchronized to the micro- processor dock, the conversion time depends on the micro- processor used. in addition, time must be allowed for reading data from the ad7878. if this task is performed during the track/ hold amplifier acquisition period, then it does not impact the overall throughput rate. however, if the read operations occur during a conversion, they may stretch the conversion time and reduce the track/hold amplifier acquisition time. the track/hold amplifier requires a minimum of 2 m s to operate to specification. the time required to read from the ad7878 depends on the number of fifo memory locations to be read and the software organization. as an example, consider an application using the adsp-2100 and the ad7878 with a throughput rate of 100 khz. the time required for the convst pulse and the adc conversion is 7.375 m s. this leaves 2.625 m s for the track/hold acquisition time and for reading the adc (both operations occurring in parallel). the adsp-2100, when operating from a 32 mhz clock, has an instruction cycle of 125 ns and an interrupt re- sponse time of 500 ns. this allows adequate time to perform 16 read operations within the time budget allowed. table ii. ad7878 throughput rate convst conversion t/h acquisition pulse width time time number of non- clock cycles 2 min 57 max applicable adsp-2100 1 250 ns min 7.125 m s max 2 m s min tms32010 2 400 ns min 11.14 m s max 2 m s min tms32020 2 400 ns min 11.14 m s max 2 m s min notes 1 adsp-2100 clock frequency = 32 mhz. 2 tms320xx clock frequency = 20 mhz. application hints good printed circuit board (pcb) layout is as important as the overall circuit design itself in achieving high speed a/d perfor- mance. the ad7878 is required to make bit decisions on an lsb size of 1.465 mv. to achieve this, the designer has to be conscious of noise both in the adc itself and in the preceding analog circuitry. switching mode power supplies are not recom- mended as the switching spikes will feed through to the com- parator, causing noisy code transitions. other concerns are ground loops and digital feedthrough from microprocessors. these factors influence any adc, and a proper pcb layout that minimizes these effects is essential for best performance. layout hints ensure that the layout for the printed circuit board has the digital and analog signal lines separated as much as possible. take care not to run any digital track alongside an analog signal track. guard (screen) the analog input with agnd.
ad7878 C13C rev. a establish a single point analog ground (star ground) separate from the logic system ground at pin 22 (agnd) or as dose as possible to the ad7878, as shown in figure 22. connect all other grounds and pin 7 (ad7878 dgnd) to this single analog ground point. do not connect any other digital grounds to this analog ground point. low impedance analog and digital power supply common returns are essential to low noise operation of the adc, so make the foil width for these tracks as wide as possible. the use of ground planes minimizes impedance paths and also guards the analog circuitry from digital noise. the circuit layouts of figures 25 and 26 have both analog and digital ground planes, which are kept separated and only joined to- gether at the ad7878 agnd pin. noise keep the input signal leads to v in and signal return leads from agnd (pin 22) as short as possible to minimize input noise coupling. in applications where this is not possible, use a shielded cable between the source and the adc. reduce the ground circuit impedance as much as possible since any poten- tial difference in grounds between the signal source and the adc appears as an error voltage in series with the input signal. figure 22. power supply grounding practice data acquisition board figure 23 shows the ad7878 in a data acquisition circuit that will interface directly to either the adsp-2100, tms32010 or the tms32020. the corresponding printed circuit board (pcb) layout and silkscreen are shown in figures 24 to 26. the only additional component required for a full data acquisi- tion system is an antialiasing filter. there is a component grid provided near the analog input on the pcb which may be used for such a filter or any other conditioning circuitry. to facilitate this option, a wire link (labelled lk1 on the pcb) is required on the analog input track. this link connects the input signal to either the component grid or directly to the buffer amplifier driving the ad7878 analog input. microprocessor connections to the pcb can be made by either of two ways: 1. 96-contact (3 row) eurocard connector. 2. 26-contact (2 row) idc connector. the 96-contact eurocard connector is directly compatible with the adsp-2100 evaluation board prototype expansion con- nector. the expansion connector on the adsp-2100 has eight decoded drip enable outputs labelled ece8 to ece1 . ece6 is used to drive the ad7878 cs input on the data acquisition board. to avoid selecting onboard ram sockets at the same time, lk6 on the adsp-2100 board must be removed. in addi- tion, the expansion connector on the adsp-2100 has four inter- rupts labelled eirq3 to eirq0 . the ad7878 alfl output connects to eirq0 . the ad7878 and adsp-2100 data lines are aligned for left justified data transfer. the 26-way idc connector contains all the necessary contacts for both the tms32010 and tms32020. there are two switches on the data acquisition board that must be set to en- able the appropriate interface configuration (see table iii). the interface connections for the tms32010/32020 and idc signal contact numbers are shown in table iv and figure 23. note the ad7878 cs input must be decoded from the address bus prior to the ad7878 evaluation board for the tms320xx interfaces. connections to the analog input (v in ) and the convst input are made via two bnc sockets labelled skt1 and skt2 on the silkscreen. if the convst input is derived from either the microprocessor or adc clock, the effects of clock noise cou- pling will be reduced. table iii. ad7878 pcb switch settings switch setting microprocessor sw1 sw2 adsp-2100 a a tms32010 b a tms32020 b b power supply connections the pcb requires two analog supplies and one 5 v digital sup- ply. connections to the analog supplies are made directly to the pcb as shown on the silk screen in figure 24. the connections are labelled v+ and vC and the range for both of these supplies is 12 v to 15 v. connection to the 5 v digital supply is made through either of the two microprocessor connectors. the +5 v and C5 v analog power supplies required by the ad7878 are generated from two voltage regulators on the v+ and vC power supply inputs (ic3 and ic4 in figure 23). component list ic1 ad711 op amp ic2 ad7878 analog-to-digital converter ic3 mc78l05 5 v regulator ic4 mc79l05 C5 v regulator ic5* 74hc00 quad nand gate ic6* 74hc04 hex inverter ic7 74hc02 quad nor gate sw1 single pole double throw sw2 double pole double throw lk1 wire link for analog input c1, c3, c5, c7, c9 10 m f capacitors c11, c13, c15 c2, c4, c6, c8, c10 0.1 m f capacitors c12, c14, c16 r1*, r2* 10 k w resistors skt1, skt2 bnc sockets skt3 26-contact (2 row) idc connector skt4 96-contact (3 row) eurocard connector *not required for adsp-2100 interface.
ad7878 C14C rev. a figure 23. data acquisition circuit using the ad7878 figure 24. pcb silkscreen for figure 23
ad7878 C15C rev. a figure 25. pcb component side layout for figure 23 figure 26. pcb solder side layout for figure 23
ad7878 C16C rev. a c1204aC1C5/97 printed in u.s.a. table iv. tms32010/tms32020 interface connections idc signal connect tms32010 tms32020 contact no. mnemonic signal signal 1r/ w r/ w 2 strb strb 3 dmrd den 4 dmwr we 5 cs cs cs 6 ready ready 7 reset reset reset 8 alfl int int 9 add0 pa0 a0 10 clk clkout clkout2 11 db10 d10 d10 12 db11 d11 d11 13 db8 d8 d8 14 db9 d9 d9 15 db6 d6 d6 16 db7 d7 d7 17 db4 d4 d4 18 db5 d5 d5 19 db2 d2 d2 20 db3 d3 d3 21 db0 d0 d0 22 db1 d1 d1 23 5 v 5 v 5 v 24 5 v 5 v 5 v 25 gnd gnd gnd 26 gnd gnd gnd outline dimensions dimensions shown in inches and (mm). 28-pin plastic dip (n-28) 28-pin cerdip (q-28) 28-pin ceramic dip (d-28) 28-terminal plcc (p-28a) 28-terminal lccc (e-28a) note 1 analog devices reserves the right to ship either cerdip or ceramic hermetic packages.

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